Chirp receiver utilizing phase processed chirp signals

ABSTRACT

A chirp receiver processes broadcast chirp signals in the frequency domain to distinguish direct path signals from multipath signals. The receiver processes received chirp signals consisting of respective pulsed frequency sweeps by combining the signals with a synchronized local chirp signal and phase adjusting and concatenating the results over multiple sweeps based on estimated clock phase errors and expected phase rotations of the direct path signals, and produces a sine wave. The phase adjustment and concatenation allows the use of longer Fast Fourier Transforms, which provide increased accuracy of frequency estimation and separate component signals that are very close in frequency. The frequency corresponding to the direct path signal is identified by the lowest frequency bin in which power is above a predetermined noise threshold. The receiver then determines a time delay based on the identified frequency and uses the time delay to calculate accurate clock phase error and position.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates generally to a ranging receiver and moreparticularly to a ranging receiver that utilizes chirp ranging signals.

2. Background Information

Ranging systems are used to determine the location or global position ofone or more objects relative to one or more transmitters. Radar systemsand Global Navigation Satellite Systems (GNSS) are two examples of wellknown ranging systems.

GNSS receivers determine their global positions based on the time delaysassociated with code and carrier signals they receive from associatedsatellites. The GNSS receivers operate in known manners to align locallygenerated versions of the codes and carriers with the received signalbased on correlation measurements. The GNSS receivers then determine thetime delay between the known transmission time of the signal and thetime of the receipt of the signal based on the phases of the local codesand carriers, and calculate pseudoranges to the respective satellitesbased on the associated time delays. A global position is determined ina known manner using the pseudoranges to three or more satellites. Agiven pseudorange is computed from the difference between the presumedtime of code transmission by the satellite and the time of receipt ofthe code at the receiver, multiplied by the speed of light. Thepseudorange value thus contains the actual physical range to thesatellite in addition to the clock errors at both the satellite andreceiver. In GNSS systems, operators of ground control networkscontinually estimate the clock drifts of the satellites and providethese data to the receivers as part of real time kinematic (RTK) orother broadcast data messages. Further, the GNSS receiver processingsoftware operating in a known manner can compute the position of thereceiver as well as the receiver clock errors, provided the receiver hassufficient numbers of measurements, and the calculated position is thuscorrected for both satellite and receiver clock errors.

The receiver receives not only line-of-sight, or direct path, satellitesignals but also multipath signals that are reflected to the receiverfrom the ground, bodies of water, nearby buildings, and so forth. Themultipath signals, which arrive at the receiver slightly later than thedirect-path signal, combine with the direct-path signal to produce adistorted received signal. The distortion of the received signaladversely affects code and, to lesser degree, carrier alignmentoperations since the correlation measurements are made using thereceived signal—including the multipath components thereof. Thedistortion may be such that the receiver attempts to align to amultipath signal instead of the direct-path signal. This is particularlytrue for multipath signals that arrive at the receiver close in time tothe receipt of the corresponding direct path signal.

One way to more accurately align the received and the locally-generatedPRN codes is to use the “narrow correlators” discussed in U.S. Pat. Nos.5,101,416; 5,390,207 and 5,495,499. It has been determined thatnarrowing the delay spacing between early and late correlationmeasurements substantially reduces the adverse effects of noise andmultipath signal distortion on the early-minus-late measurements. Thedelay spacing is narrowed such that the noise correlates in the earlyand late correlation measurements. Also, the narrow correlators areessentially spaced closer to a correlation peak that is associated withthe punctual PRN code correlation measurements than the contributions ofmany of the multipath signals. Accordingly, the early-minus-latecorrelation measurements made by these correlators are significantlyless distorted than they would be if they were made at a greaterinterval around the peak.

Another way to more accurately align the received and thelocally-generated PRN codes is to use a multipath mitigation processingtechnique that iteratively produces estimates of the direct path signaland one or more of the multipath signals. One such technique isdescribed in U.S. Pat. Nos. 5,615,232 and 6,692,008. Another techniquethat uses multiple correlators is described in U.S. Pat. No. 5,414,729.Yet another multipath mitigation technique is described in U.S. Pat. No.7,738,536.

Note that all GNSS methods of multipath mitigation are limited by thebroadcast bandwidth of these systems. The limit of the GNSS multipathmitigation techniques to separate a multipath signal from a direct pathsignal utilizing a 20 MHz broadcast bandwith and signal processing isabout 4 meters. In other words, if the multipath signal overlap of thedirect path signal is within 4 meters, the mitigation techniques cannotclearly distinguish the direct path signal from the combined signal anda corrupted tracking error may result. It is well known that the use ofwider band systems, such as Ultra Wide Band systems that have much widerbandwidths then GNSS, can support multipath mitigation techniques thatcan discern the difference between the direct signal and the multipathsignal when the two are closer together. For example, a system utilizinga 6 GHz Ultra Wide Band signal should theoretically be 300 times moreaccurate than one utilizing a 20 MHz GNSS signal.

In certain systems pseudolites are utilized to provide additionalranging signals, particularly in environments in which the pseudolitescan be placed to essentially avoid certain reflectors, such asparticular buildings and so forth, and/or in environments in whichportions of the view of the sky may be blocked by buildings and soforth. The pseudolites are ground-based transmitters that transmitranging signals, such as GNSS-like signals containing PRN codes. Thepseudolite signals, like the GNSS signals, are reflected from reflectorsthat are nearby the antenna, such as the ground, the antenna frame andso forth, and thus, multipath mitigation techniques may be required forthe pseudolite signals as well.

The multipath techniques described above work well, and the systems canobtain centimeter accuracies for clock phase measurements inenvironments in which the multipath signals arrive relatively close intime to the direct path signals, i.e., the multipath signal and thedirect path signal are separated by about 4 meters. However, multipathsignals that are closer than 4 meters to the direct path signal, thatis, multipath signals that received within nanoseconds of the directpath signal, continue to be sources of error. Environments in which sucherrors may occur are, for example, construction sites in which a GNSSreceiver may be in use in an excavation cavity with contours that act asnearby signal reflectors for both GNSS satellite signals and pseudolitesignals.

Accordingly, there remains a need for a ranging receiver that canprovide for even greater accuracy in situations in which multipathsignals arrive at the receiver antenna particularly close in time to thedirect path ranging signals.

SUMMARY OF THE INVENTION

A chirp receiver processes broadcast chirp signals in the frequencydomain using a Fast Fourier Transform (FFT) to distinguish the directpath signal from the respective multipath signals. The chirp receiverprocesses the received chirp signals, which consist of respective pulsedfrequency sweeps, by combining a received chirp with a synchronizedlocally generated chirp and phase adjusting and concatenating theresults over multiple chirps to produce sine waves. The phase adjustmentand concatenation, which is, based on the estimated clock phasedifferences between the receiver clock and the received chirps, allowsfor a longer length, and therefore narrower band, FFT that produces highfidelity frequency measurements that distinguish the various signalscontained in the composite received signal. The phase adjustment andconcatenation thus allows for the separation of multipath signals thatare in very close proximity to the direct path signal, with up to a 1millimeter accuracy.

The frequency corresponding to the direct path signal is identified bythe lowest frequency bin in which power is above a predetermined noisethreshold. The receiver then determines measurements of clock phasedifferences between the local clock and the received signal based on theidentified frequency. The clock phase differences may be used tocalculate the pseudorange to a chirp signal transmitter that is at aknown location and using a known clock and frequency source.Alternatively, or in addition, the clock phase differences may be usedto transfer accurate time and frequency across a wireless link.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention can be better understood with reference to theaccompanying drawings, of which:

FIG. 1 is a functional block diagram of a system that utilizes a chirpreceiver;

FIG. 2 is an illustration of a chirp signal transmitted by a chirptransmitter of FIG. 1;

FIG. 3 is a more detailed functional block diagram of the chirp receiverof FIG. 1;

FIG. 4 illustrates phase precessing performed by the chirp receiver ofFIG. 1; and

FIG. 5 is a more detailed functional block diagram of a chirp generatorof the receiver of FIG. 3.

DETAILED DESCRIPTION OF AN ILLUSTRATIVE EMBODIMENT

For ease of explanation, the operations of the system are describedfirst with the co-located GNSS and chirp receivers. The operations arethen described for a chirp receiver operating without the co-locatedGNSS receiver.

Referring to FIG. 1, a ranging system 100 includes one or more chirptransmitters 102 in known locations, a chirp receiver 104 and a GNSSreceiver 106 are co-located on a vehicle 108. Additional GNSS receivers(not shown) may be associated with one or more of the chirp signaltransmitters, to provide both time synchronization and the locations ofthe transmitters based on ranging signals transmitted by GNSSsatellites. Alternatively, the chirp transmitters 102 may be placed inpredetermined locations or be provided with their time and locations andtime synchronization signals at the time of placement using GNSSreceivers or other devices to determine the time and locations, andthus, GNSS receivers need not be co-located with the chirp transmitters.

However, without the co-locating a method of providing clocksynchronization is needed. One method of providing the clocksynchronization is to monitor respective free running chirp transmitters102 using a chirp receiver that is connected to a known clock andfrequency source, and then broadcast transmitter clock offset and chirprate terms to the chirp receiver 108 over a communication link. Thismethod is similar to GNSS RTK in which a base GNSS receiver at a knownlocation broadcasts information relating to GNSS satellite clock drift,and the GNSS receivers utilize the information in a well known manner toachieve centimeter accuracy. A convenient place to monitor the chirptransmitter clocks, as well as the GNSS satellite clocks, is in acombined GNSS and chirp RTK reference receiver 112.

The chirp receiver 104 includes an antenna 105 for receiving the chirpsignals broadcast by the chirp transmitters 102. The GNSS receiver 106includes an antenna 107 for receiving the GNSS satellite signals. Asshown, one or more reflectors 110 _(i), such as a hill 110 ₁ and theground 110 ₂, reflect the ranging signals (solid line) as multipathsignals (dotted lines) to the antennas 105 and 107 of the chirp receiver104 and the GNSS receiver 106, respectively. The GNSS receiver 106operates in a known manner to determine an estimated position of thevehicle 108, using the signals transmitted by GNSS satellites 120 and asappropriate the GNSS RTK information. The chirp receiver 104 operates,as discussed below with reference to FIGS. 3-5, to determine veryaccurate clock phase differences between the received chirp signal and alocal clock, and thus, the associated time delay. The chirp antenna 105and GNSS antenna 107 may be co-located. If separated on the vehicle 108,care must be taken to account for the orientation and separationdifference between the antennas when computing a position based onmeasurements from both GNSS and chirp signals.

Referring now also to FIG. 2, the chirp transmitters 102, atpredetermined transmission times, transmit ultra wide band chirp radiofrequency (RF) signals. One form of which is shown as curve 200. In theexample, the predetermined times are based on GPS time, though thepredetermined times may be based on any distributed clock. Each chirptransmitter 102 is assigned a predetermined time slot, or time offset,from the GPS or other transmission time, so that the chirp receiver 104can distinguish the signals from the respective transmitters.

A chirp RF signal is a swept frequency signal that is generated in apulsed fashion. Each pulse consists of one or more frequency sweepsacross the operating bandwidth. In some cases, pulse-shaping may beapplied to the pulses before transmission, to reduce spectral emissions.The direction of the sweep can be up or down or, for multiple sweeps inthe same pulse, an arbitrary mixture of up and down.

The manner in which the frequency changes with time can follow differentfunction shapes such as linear, arc tangent, logarithmic etc. FIG. 2illustrates a linear sweep. If a sweep other than a linear sweep isutilized in the system 100, the inverse of the function is applied tothe received chirp signal, in order to produce an end-to-end linearsweep. The advantage of an end-to-end linear sweep is that a delay in areceived signal translates directly to a frequency and a phase shift.

A transmitted chirp signal x(t) is represented by the followingequation:x(t)=cos(2π∫₀ ^(t) f(τ)dτ)  Eqn. 1for a linear chirp,

$\begin{matrix}{{f(\tau)} = {{f_{0} + {\frac{\Delta\; f}{T_{sweep}}\tau}} = {f_{0} + {k\;\tau}}}} & {{Eqn}.\mspace{14mu} 2}\end{matrix}$Substituting eqn. 2 into eqn. 1 and performing the integration:

$\begin{matrix}{{x(t)} = {{\cos( {2\;{\pi( {f_{0} + {\frac{k}{2}t}} )}t} )} = {{\cos( {\phi(t)} )} = \frac{( {{\mathbb{e}}^{j\;{\phi{(t)}}} + {\mathbb{e}}^{{- j}\;{\phi{(t)}}}} )}{2}}}} & {{Eqn}.\mspace{14mu} 3}\end{matrix}$

To determine the time delay of a received chirp signal, the receivedchirp signal is combined with a locally-generated single-sidebandversion of the chirp signal:

$\begin{matrix}\begin{matrix}{{y(t)} = {{x( {t - t_{d}} )}2\;{\mathbb{e}}^{j\;{\phi{(t)}}}}} \\{= {\frac{( {{\mathbb{e}}^{j\;{\phi{({t - t_{d}})}}} + {\mathbb{e}}^{{- j}\;{\phi{({t - t_{d}})}}}} )}{2}2\;{\mathbb{e}}^{j\;{\phi{(t)}}}}} \\{= {{\mathbb{e}}^{j{({{\phi{(t)}} + {\phi{({t - t_{d}})}}})}} + {\mathbb{e}}^{j{({{\phi{(t)}} - {\phi{({t - t_{d}})}}})}}}}\end{matrix} & {{Eqn}.\mspace{14mu} 4}\end{matrix}$After low pass filtering, the combined signal y(t) is represented by thefollowing equation:

$\begin{matrix}{{y(t)} = {{\mathbb{e}}^{j{({{\phi{(t)}} - {\phi{({t - t_{d}})}}})}} = {{\mathbb{e}}^{j{({{{kt}_{d}f_{0}t_{d}} - {\frac{k}{2}t_{d}^{2}}})}} \approx {\mathbb{e}}^{j{({{{kt}_{d}t} + {f_{0}t_{d}}})}}}}} & {{Eqn}.\mspace{14mu} 5}\end{matrix}$where t_(d) is the delay between signal transmission and receipt, andthe subtracted t_(d) ² term is, for typical values of t_(d), negligible.

Note that Eqn. 5 contains a constant frequency defined by kt_(d) and aconstant phase shift defined by f₀t_(d). Accordingly, both the phase andthe frequency of the received signal convey information about the timedelay t_(d) between the transmission of the signal and the receipt ofthe signal. As discussed in more detail below, the frequency is utilizedin the system 100 to determine the time delay of the direct path signalcomponent of the received chirp signal and the phase is used toconcatenate the signal samples for processing.

Referring now to FIGS. 1 and 3, the chip receiver 104 includes an RFfront end 300 that processes the received chirps, which consist of thedirect path chirp signal and corresponding multipath signals and is thusa composite signal provided by the antenna 105. The receiver 104 firstamplifies the received signal in a low noise amplifier (LNA) 302, inorder to reduce the adverse effects of the noise contributed by thedown-stream components. The LNA 302 has a sufficiently high third orderintercept point (IP3), to accommodate the power of the receivedcomposite signal without introducing significant distortion.

The chirp signal is next provided to a complex mixer 304 that mixes thesignal with a locally generated chirp signal provided by a chirpgenerator 306, which is discussed in more detail below with reference toFIG. 5.

After mixing, the complex intermediate frequency (IF) signal is providedto a low pass filter 312 and amplifier 314 which essentially attenuatepower above the frequencies of interest and bring the signal to a levelcompatible with the operation of an Analog to Digital converter (ADC)316. The ADC provides digital samples of the complex IF signal to adigital signal processor (DSP) 400. As appropriate, the low pass filterand amplifier may operate in a known manner to provide variable gain, toaccommodate an ADC with a lower dynamic range that may otherwiseadversely affect the conversion of the signals at the extremes of theincoming signal power.

The various operations of the DSP 400 are discussed below. For ease ofexplanation, the operations are referred to and referenced in thedrawing as processing blocks.

The sample signals provided by the ADC 316, which are time domainsignals, are supplied to a time domain to frequency domain conversionblock 402, which operates in a known manner to provide equivalentsignals for processing in the frequency domain.

A phase precessor block 408 adjusts the phases of the samplescorresponding to respective chirps in accordance with phase informationprovided by a phase calculator 410, to essentially enforce a continuityof the pulsed signal being tracked. The phase precessor block adjuststhe starting phases of the samples corresponding to respective chirps insmall phase increments so that a continuous sinusoidal signal isproduced during the course of a measurement epoch and the edges ofadjacent chirps align. This is referred to hereinafter also as “phasestitching.” An example is illustrated in FIG. 4, in which samplescorresponding to chirps 2 and 3 are phase adjusted by π/2 and πrespectively, and are then concatenated with the samples of chirp 1, toform a sine wave 4000. The phase adjusted and concatenated signals arethen provided to an FFT block 416 for processing.

Briefly, the phase precessor 408 adjusts the phase of, for example, asample taken at ½ cycle of a second chirp to a phase corresponding to 1½cycles of the first chirp, and the chirp signal samples are thenconcatenated and appear to an FFT block 416 as continuous sweeps orsinusoidal cycles. The phase calculator 410 determines the adjustmentsbased on the expected phase of the direct path signal, which is, inturn, based on phase information associated with an estimated range or,in the example with co-location with a GNSS receiver, the known GNSSposition of the chirp receiver, as well as clock phase error estimatesprovided by the FFT block 416. As appropriate, movement or trajectoryinformation from inertial or other movement sensors (not shown) are alsoutilized. The operation of the phase calculator 410 is discussed in moredetail below.

The phase precessed signals are held in a buffer 414 that allows thesignal samples to be concatenated, or phase stitched, over multiplechirps in order to create a sufficient number of signal samples that canbe processed as if they are continuous for a longer length. It is wellknown that an FFT will produce higher fidelity frequency measurementswith longer length data sample sets. A swinging buffer may be used toallow samples to be loaded at one end of the buffer while the FFTprocessing block 416 takes data from an opposite end of the buffer. Thelength of the buffer is a design choice. However, the length isconstrained by the stability of an oscillator 308 and the receiver clock(not shown), since both must be stable over the measurement interval togreater than the accuracy requirement of the measurements.

The FFT block 416 is essentially a bank of matched filters thatcorrelate frequency components of the data to a set of sinusoidal basisfunctions. A large FFT is used to provide fine frequency resolutionassociated with the measurement accuracy. It is possible to replace theFFT block 416 with a bank of correlators that operate around thefrequency associated with the line-of-sight component of the receivedsignal. As discussed below, the associated frequency is based on theestimated chirp propagation distance, that is, the estimated range,estimated phase offsets, and estimated clock phase errors. The advantageof the FFT block 416 is, however, that the processing provides a betterobservation of the overall multipath environment and also facilitatescoarser quantization of the start times of the locally generated chirps.Further, the processor is easy to implement in an off-the-shelf FieldProgrammable Gate Array (FPGA).

The FFT is performed on the phase adjusted and concatenated, or phasestitched, signals to estimate the frequency most closely associated withthe time delay of the received chirp signal, and determine the clockphase errors between the received chirp signal and the receiver clock.The clock phase errors may be used in the calculations of thepseudorange to the chirp transmitter 102, as discussed below.Alternatively, or in addition, the clock phase errors may be used toprovide accurate time over a wireless link.

The chirp signal transmitter 102 is restricted to particular powerlevels by government regulation, to avoid interference between the ultrawide band chirp signals and the signals transmitted by other nearbytransmitters. Accordingly, the chirp receiver 104 averages manymeasurements, after phase adjustment and concatenation, to producesignals with sufficient power for the FFT processing.

Also, because of power regulations, the broadcast chirp signals aresparse, i.e., there are relatively long periods of time between thetransmitted chirp pulses. Accordingly, the phase adjustment andconcatenation, or phase stitching, of the signal samples of therespective chirps are based on the calculated phase errors between thereceived chirp signal and the locally-generated chirp signal, which isdriven into synchronism with the received direct path signal. If thelocal generator is not in synchronism, the phase stitching is based onthe expected phase differences corresponding to the estimated range andclock offset estimates. The frequencies and phases of the multipathcomponents of the received signal differ from those of the direct pathcomponent, because the multipath signals arrive later in time that thedirect path signals. Accordingly, the components of the multipathsignals phase stitch and average sub-optimally.

The FFT of the phase stitched and averaged signals in FFT block 416 thusresults in relatively high power in the bin corresponding to thefrequency associated with the time delay of the direct path componentand lower power in the frequency bins that correspond to the delays ofthe respective multipath signals. The frequency bins may correspond todelay times equal to ½ millimeter of distance. Since the multipathsignals always arrive later than the direct path signals, the directpath signal is discernable in the lowest frequency bin in which thepower is above a predetermined minimum threshold associated with noise.Notably, the bin associated with the direct path signal should also havesignificantly higher power than the adjacent higher frequency bins,which are associated with the multipath signals. In cases where thedelay between the receipt of the direct path signal and the multipathsignal is very small, interpolation may be performed between adjacentbins that each have substantial power, in order to increase resolution.

As discussed, the phase calculator 410 determines a phase adjustment orprecession for the samples of the respective chirps in order to phasestitch the chirp samples into sine waves for use in the FFT processingblock 416. The chirps 1, 2 and 3 (FIG. 4) are separated in time and thephase calculator determines the expected phase rotations occurringbetween the respective chirps in order to correctly phase stitch thesignal samples from the respective chirps into a continuous sine wave.Accordingly, once the local chirp source is grossly aligned with thereceived chirp signals, that is, they at least overlap, the receiver maycalculate the phase rotations or utilize a look up table ofpre-calculated values, to adjust the phases for the phase stitching ofthe chirps. The phase adjustment values are calculated based on theknown chirp rate and the length of the sweep, the start and endfrequencies of the sweep, and as appropriate the estimated range and theestimated clock phase errors as determined by the FFT block 416. Asnecessary, the calculations include the sensed movements of the vehicle108, as measured by local inertial or other sensors (not shown).

More specifically, coarse acquisition can be achieved by using alow-resolution FFT with a time duration approximately equal to the sweepduration. The start time of the receiver chirp is adjusted until anacceptable frequency is detected by the FFT. Also, as long as thereceiver is not moving, multiple samples can be coherently summed toimprove the signal to noise ratio. Once an overlap between the receivedchirp and the locally generated chirp is determined, a coarse frequencyand phase can then be extracted from the FFT.

Once this is done, the FFT resolution can be doubled by adjusting thephase of the next sweep, in the example, a second sweep, so that thesecond sweep can be concatenated with the first sweep to form the longerduration time sequence required for the higher resolution FFT. With nomovement, the phase adjustment to concatenate the sweeps is calculatedfrom the length and frequency of the sweep. The phase adjustment is thephase increment required to make the starting phase of the second sweepthe same as the ending phase of the first sweep, to produce a continuoussinusoidal signal. As additional sweeps are concatenated into thesinusoidal signal, the resolution of the FFT can increase, and betterfrequency resolution and phase can be determined. This process can berepeated until the necessary resolution is obtained.

The estimated range, or pseudorange, along with the estimated clockoffsets could assist the initial acquisition by predicting when thechirps will arrive at the receiver, and the search space can thus benarrowed. Without the estimations, the receiver must test all possiblephase offsets with respect to the transmit pattern in order to determinean overlap between the received chirp and the locally generated chirp.After acquisition and in steady state tracking, the numericallycontrolled oscillator (“NCO”) 308 of the chirp tracking loop 422estimates the pseudorange directly so the theoretical or estimated rangeis not as useful.

If the estimated range has a rate to it, that is, if the receiver ismoving or one of the clocks is drifting, the estimated pseudorange andpseudorange-rate could both be used to help narrow the search space.

The system may also utilize a table of pre-calculated phase adjustmentsassociated with various clock phase errors. The table is entered usingthe identified frequency, and the phase calculator utilizes the valuesfrom the table along with the expected phase adjustment associated withthe estimate range and the predetermined clock offset, to determine thephase adjustments for the signal samples of the respective chirps. Ifthe receiver is stationary, the table may include in the pre-calculatedvalue the phase adjustment associated with the path that the directsignal travels.

Alternatively, the phase calculator 410 may utilize early and late FFTsin addition to the on-time or punctual FFT 416 in a tracking loop. Theearly FFT adjusts the phase of the samples by, for example 45°, thepunctual FFT by 90° and the late FFT by 270°, and the FFTs form atracking loop that tracks the phase rotations. Thus, the phasecalculator 410 adjusts the rotations for the early, punctual and lateFFT's to drive the punctual FFT toward a predetermined offset frequency.The differences in phase offsets between the FFTs are narrowed as powerin the punctual FFT is driven closer to the offset frequency.

The results of the early, punctual, and late power differences are thusused to adjust the “phase stitching” of the wavelets corresponding tothe chirp pulses. If more power is seen in the Early FFT estimate, the“phase stitching” tracking loop reduces the amount of phase rotationapplied to the chirp signal samples. Likewise, if there is more powerreceived in the Late FFT estimate, then the “phase stitching” trackingloop increases the amount of phase rotation applied to the chirp signalsamples, and so forth, until the Punctual FFT contains the highest powervalue and the Early and Late FFT power values about the same.

Once the FFT tracking loop is synchronized to the direct path signal,each chirp should stitch perfectly with the previous and next chirps toform a sine wave associated with the frequency of the direct pathsignal. This “stitching correction” is only optimal for a very narrowband of frequencies. Other frequencies received in the signal, forexample, from multipath signals, will not stitch together as optimallyas the direct path signal and so will have reduced power readings fromthe FFT.

The results of the FFT processing or, as appropriate, the punctual FFTare provided to a frequency error-to-time delay error translator block418, which drives a rate block 310 that, in turn, adjusts the NCO 308 todrive the local chirp generator 306 into synchronism with the receivedchirp signal. Alternatively, the phase of the locally generated chirpmay be shifted to realign the FFT output instead of adjusting the chirprate.

The frequency error-to-time delay error block 418 is thus part of achirp tracking loop 422 that adjusts the local chirps based on the FFTmeasured frequency associated with the direct path signal. To take thebest advantage of the FFT processing 416, a predetermined offset is usedbetween the local and the received chirps, such that the result of theFFT will be the predetermined offset frequency when the local chirpgenerator is synchronized to the received chirps. However, producing theIF signal with a local chirp that is offset by the predetermined amountresults in a small loss of signal at the end of the sweep, due to thefact that the transmitter completes the sweep before the receiver does.

The receiver's chirp may instead be offset in frequency by apredetermined amount such that the two chirps overlap completely whentracking, with no loss of power yet providing an offset frequency. If anobserved frequency shift occurs, for example, a lower frequency than theoffset frequency is observed, the locally generated pulses are sloweddown slightly by the NCO 308 to re-align the frequency to the offsetfrequency. Conversely, if a higher frequency is observed in the FFToutput, the locally generated chirps are sped up by the NCO 308 torealign the FFT output frequency to the offset frequency.

When the FFT processing indicates that the chirp generator 306 is insynchronism with the received chirp signal, the NCO represents thebroadcast phase of the received chirp signal at any point in time.Accordingly, phase measurements of the NCO are taken periodically, forexample, every 1 second in synchronism with a local 1PPS strobeassociated with the receiver's time clock. The differences between theNCO phase measurements and the local time, e.g., the 1 PPS in theexample, and are then multiplied by the speed of light to determinepseudoranges to the chirp transmitter, in order to further deriveposition, time or clock offset information.

If the approximate clock phase offset between the transmitter and thereceiver cannot be estimated at start-up, if for example there is noGNSS receiver 106 at the location of the chirp receiver 104 and theposition of the receiver is not yet known, a search procedure as is wellknown in the art is necessary to grossly align the received chirp withthe locally generated chirp. The receiver thus varies the rate of thechirp generator and utilizes the results of the FFT to determine whenthe local chirp at least overlaps with the received chirp based on theFFT power. Once the received chirps and locally generated chirps have atleast some overlap, FFT processing 416 is used to more accurately bringthe local chirp generator into synchronization with the received chirpsas discussed above.

In addition to the above-described components, the DSP 400 may alsoimplement an optional impairment correction block 404, to correct forgroup delay distortion of input filters, amplifiers and other RFcomponents, and/or amplitude distortion of the amplifiers and other RFcomponents and/or an I/Q imbalance, if I/Q mixing is done in the analogdomain. The correction values may, for example, be determined duringcalibration operations at the manufacturer and stored in the DSP.

The DSP 400 may also implement an optional interference blanking block406, to isolate the frequencies that have higher than expected powerlevels. To prevent the samples from corrupting the measurements, theblock either blanks the corresponding samples or substitutes aninterpolated waveform for the samples.

A decimator block 412 operates in a known manner to reduce the samplingrate while preventing aliasing of noise and other higher frequencyinterference from corrupting the signal in the desired bandwidth. Inaddition, while the phase precessing is shown by block 408, theoperation may instead occur at one or more other locations in theprocessing and may include a phase precessing component in the RFprocessing.

Advantageously, a chirp receiver operating as discussed above is able todetermine the time delay for a direct path component of the receivedchirp signal in the frequency domain by processing multiple chirps,phase stitching the results based on the frequency and/or phasedifferences as measured between the received chirp and the receiverclock and performing an FFT. The receiver can then precisely determinethe phase relationship between the transmitted chirp and the receiverclock based on the calculated the time delays.

Referring now to FIG. 5, the chirp source 306 is illustrated in moredetail. A direct digital synthesis (DSS) device 500 generates areference signal which is then multiplied up in frequency by a wide bandphase lock loop 518 to drive a voltage controlled oscillator (VCO) 516,which produces the chirp, or sweep, signal while maintaining accuracy inthe generated frequencies and sweep trajectories. In the example, thereference signal sweeps between 88 MHz and 160 MHz and the PLL 518multiplies the sweeps up in frequency by 64 to 3.1 to 10.6 GHz. Thechirp generator 306 further includes a digital to analog converter (DAC)502, a smoothing filter 504 and an amplifier 508 that together conditionthe signal produced by the DSS 500 before the signal is mixed with theoutput of the PLL 518. The PLL includes a loop filter 514 that smoothesthe results and the VCO 516, which responds to the signal from the loopfilter and produces the desired frequency sweep. Alternatively, otherknown devices may be utilized to produce the chirp signal.

An optimal phase compensation block 506 may be included between thefilter 504 and the amplifier 508, such that the phase of the chirp canbe adjusted to drive the local chirps to synchronism with the receivedchirps.

The chirp source 306 may also include delays (not shown) to generateearly, on-time/punctual and late versions of the chirp signal, asappropriate.

The RF front end 300 of the chirp receiver 104 may instead utilize abaseband signal and analog I/Q mixing, which requires separate I and QADCs, filters and gain components per channel.

While a single channel is shown, multiple channels may be utilized toprocess signals received from multiple chirp signal transmitters 102.Alternatively, the multiple channels may be utilized to speed up theprocessing of the signals received from a single chirp signaltransmitter.

The foregoing description has been directed to specific embodiments ofthe invention. It will be apparent, however, that other variations andmodifications may be made to the described embodiments, with theattainment of some or all of the advantages of such. For example, theprocessing blocks described as operations performed by a DSP may beperformed by one or more processors, by software, hardware, firmware orcombinations thereof. An additional phase precessor may operate onanalog signals or be included in the chirp signal source and operate inconjunction with the DSS, to provide continuity with the digital phaseprecessor block operating also to further align the sweep samples.Accordingly, this description is to be taken only by way of example andnot to otherwise limit the scope of the invention. Therefore, it is theobject of the appended claims to cover all such variations andmodifications as come within the true spirit and scope of the invention.

What is claimed is:
 1. A method of measuring clock phase errorcomprising: receiving, at a chirp receiver, a chirp signal consisting ofa plurality of chirps that are pulsed frequency sweeps transmitted by achirp transmitter; mixing the received chirp signal with a locallygenerated chirp signal that is driven into synchronism with the receivedchirp signal; adjusting samples of the mixed signals corresponding tomultiple sweeps based on estimated clock phases and expected phaserotations of direct path signal components of the respective receivedchirps to align edges of adjacent chirps and concatenating the samplesto produce a sinusoidal signal over a measurement epoch; processing theconcatenated samples in a frequency domain and determining a frequencycorresponding to a direct path signal component as the lowest frequencyassociated with power above a predetermined threshold; calculating atime delay associated with the lowest frequency determined for thedirect path signal component and determining a clock phase error betweena receiver clock and the received signal.
 2. The method of claim 1,wherein the locally generated chirp signal is generated by a directdigital synthesis device.
 3. The method of claim 1, wherein processingin the frequency domain consists of a Fast Fourier Transform (FFT). 4.The method of claim 3 further including altering starting phases of thesamples from respective sweeps to provide a continuous sinusoidal signalfor FFT processing.
 5. The method of claim 4, wherein the time delay ofthe direct path signal component is determined by interpolating betweenfrequencies associated with multiple bins having power above apredetermined threshold.
 6. The method of claim 1, wherein the adjustingand concatenation occur by adjusting the phase of the samples based onexpected phase rotations between the sweeps, the associated estimatedclock phase errors, and propagation delay and rate of change in delaybased on an estimated position of the receiver and sensed movement ofthe receiver.
 7. The method of claim 1 further including determining areceiver clock offset from a transmitter clock based on the clock phaseerror.
 8. The method of claim 7 further including determining a range tothe receiver based on a transit time and the clock phase error.
 9. Themethod of claim 1 further including computing position and clockcomponents of the receiver with measurements from chirp pseudorangemeasurements.
 10. The method of claim 9 further including a GlobalNavigation Satellite System (GNSS) receiver co-located with the chirpreceiver to determine a global position using signals from a pluralityof GNSS satellites.
 11. The method of claim 10 wherein the position andclock components of the chirp receiver are computed with measurementsfrom the GNSS receiver and chirp pseudorange measurements.
 12. A chirpintercept receiver comprising: an antenna that receives a chirp signalconsisting of pulsed frequency sweeps transmitted from a chirp signaltransmitter at a known location; a chirp signal source that generates alocal version of the chirp signal; a mixer that mixes the received andlocal versions of the chip signal; and at least one processor configuredto: adjust phase samples of the mixed signal over multiple sweeps basedon estimated clock phases and expected phase rotations of direct pathsignal components of respective received chirps to align edges ofadjacent chirps and concatenate the samples to produce a sinusoidalsignal over a measurement epoch, process the concatenated samples in afrequency domain to determine a frequency associated with a directsignal component of the received chirp signal as the lowest frequencywith power above a predetermined threshold, and calculate a phasedifference between the received chirp signal and a receiver clock basedon a time delay associated with the lowest frequency determined for thedirect path chirp signal component.
 13. The receiver of claim 12,wherein the at least one processor utilizes phase rotation associatedwith receiver and transmitter clock phase errors, propagation delay andreceiver movement.
 14. The receiver of claim 12, wherein the receiverprocesses the adjusted and concatenated samples using a Fast FourierTransform (FFT) with frequency bins associated with phase offset betweenreceived pulse and local clock of the receiver.
 15. The receiver ofclaim 14, wherein the receiver performs FFT associated with early,punctual and late versions of phase adjusting estimates.
 16. Thereceiver of claim 12 further includes a Global Navigation SatelliteSystem (GNSS) receiver co-located with the chirp receiver to determine aglobal position using signals from a plurality of GNSS satellites. 17.The method of claim 16 further includes position and clock components ofthe chirp receiver are computed with measurements from the GNSS receiverand chirp pseudorange measurements.